Communications  connectors including transmission lines having impedance discontinuities that improve return loss and/or insertion loss performance and related methods

ABSTRACT

Communications plugs are provided that include a housing that receives the conductors of the communication cable. A printed circuit board is mounted at least partially within the housing. A plurality of plug contacts are on the printed circuit board, and the printed circuit board includes a plurality of conductive paths that electrically connect respective ones of the conductors to respective ones of the plug contacts. First and second of the conductive paths are arranged as a first differential pair of conductive paths that comprise a portion of a first differential transmission line through the communications plug, where the first differential transmission line includes a first transition region where the impedance of the first differential transmission line changes by at least 20% and a second transition region impedance of the first differential transmission line changes by at least 20%.

CROSS REFERENCE TO RELATED APPLICATION

The present application claims priority under 35 U.S.C. §120 to U.S.application Ser. No. 14/090,073, filed Nov. 26, 2013, which in turnclaims priority under 35 U.S.C. §119 to U.S. Provisional ApplicationSer. No. 61/755,581, filed Jan. 23, 2013, the entire content of each ofwhich is incorporated herein in its entirety as if set forth fullyherein.

FIELD OF THE INVENTION

The present invention relates generally to communications connectorsand, more particularly, to communications connectors such as modularplugs that may exhibit improved return loss and/or insertion lossperformance.

BACKGROUND

Many hardwired communications systems use plug and jack connectors toconnect a communications cable to another communications cable or to apiece of equipment such as a computer, printer, server, switch or patchpanel. By way of example, high speed communications systems routinelyuse such plug and jack connectors to connect computers, printers andother devices to local area networks and/or to external networks such asthe Internet. FIG. 1 depicts a highly simplified example of such ahardwired high speed communications system that illustrates how plug andjack connectors may be used to interconnect a computer 11 to, forexample, a network server 20.

As shown in FIG. 1, the computer 11 is connected by a cable 12 to acommunications jack 15 that is mounted in a wall plate 19. The cable 12is a patch cord that includes a communications plug 13, 14 at each endthereof. Typically, the cable 12 includes eight insulated conductors. Asshown in FIG. 1, plug 14 is inserted into an opening or “plug aperture”16 in the front side of the communications jack 15 so that the contactsor “plug blades” of communications plug 14 mate with respective contactsof the communications jack 15. If the cable 12 includes eightconductors, the communications plug 14 and the communications jack 15will typically each have eight contacts. The communications jack 15includes a wire connection assembly 17 at the back end thereof thatreceives a plurality of conductors (e.g., eight) from a second cable 18that are individually pressed into slots in the wire connection assembly17 to establish mechanical and electrical connections between eachconductor of the second cable 18 and a respective one of a plurality ofconductive paths through the communications jack 15. The other end ofthe second cable 18 is connected to a network server 20 which may belocated, for example, in a telecommunications closet of a commercialoffice building. Communications plug 13 similarly is inserted into theplug aperture of a second communications jack (not pictured in FIG. 1)that is provided in the back of the computer 11. Thus, the patch cord12, the cable 18 and the communications jack 15 provide a plurality ofelectrical paths between the computer 11 and the network server 20.These electrical paths may be used to communicate electrical informationsignals between the computer 11 and the network server 20.

When signals are transmitted over a conductor (e.g., an insulated copperwire) in a communications cable, electrical noise from external sourcesmay be picked up by the conductor, degrading the quality of the signal.In order to counteract such noise sources, the information signals inthe above-described communications systems are typically transmittedbetween devices over a pair of conductors (hereinafter a “differentialpair” or simply a “pair”) rather than over a single conductor. The twoconductors of each differential pair are twisted tightly together in thecommunications cables and patch cords so that the eight conductors arearranged as four twisted differential pairs of conductors. The signalstransmitted on each conductor of a differential pair have equalmagnitudes, but opposite phases, and the information signal is embeddedas the voltage difference between the signals carried on the twoconductors of the pair. When the signal is transmitted over a twisteddifferential pair of conductors, each conductor in the differential pairoften picks up approximately the same amount of noise from theseexternal sources. Because approximately an equal amount of noise isadded to the signals carried by both conductors of the twisteddifferential pair, the information signal is typically not disturbed, asthe information signal is extracted by taking the difference of thesignals carried on the two conductors of the differential pair, and thissubtraction process may mostly cancel out the noise signal.

Referring again to FIG. 1, it can be seen that a series of plugs, jacksand cable segments connect the computer 11 to the server 20. Each plug,jack and cable segment includes four differential pairs, and thus atotal of four differential transmission lines are provided between thecomputer 11 and the server 20 that may be used to carry two waycommunications therebetween (e.g., two of the differential pairs may beused to carry signals from the computer 11 to the server 20, while theother two may be used to carry signals from the server 20 to thecomputer 11). Unfortunately, the proximities of the conductors andcontacting structures within each plug-jack connection (e.g., where plug14 mates with jack 15) can produce capacitive and/or inductivecouplings. These capacitive and inductive couplings in the connectors(and similar couplings that may arise in the cabling) give rise toanother type of noise that is known as “crosstalk.”

In particular, “crosstalk” refers to unwanted signal energy that iscapacitively and/or inductively coupled onto the conductors of a first“victim” differential pair from a signal that is transmitted over asecond “disturbing” differential pair. The induced crosstalk may includeboth near-end crosstalk (NEXT), which is the crosstalk measured at aninput location corresponding to a source at the same location (i.e.,crosstalk whose induced voltage signal travels in an opposite directionto that of an originating, disturbing signal in a different path), andfar-end crosstalk (FEXT), which is the crosstalk measured at the outputlocation corresponding to a source at the input location (i.e.,crosstalk whose signal travels in the same direction as the disturbingsignal in the different path). Both types of crosstalk comprise anundesirable noise signal that interferes with the information signalthat is transmitted over the victim differential pair.

While methods are available that can significantly reduce the effects ofcrosstalk within communications cable segments, the communicationsconnector configurations that were adopted years ago—and which still arein effect in order to maintain backwards compatibility—generally did notmaintain the arrangement and geometry of the conductors of eachdifferential pair so as to minimize the crosstalk coupling between thedifferential pairs in the connector hardware. For example, pursuant tothe ANSI/TIA-568-C.2 standard approved Aug. 11, 2009 by theTelecommunications Industry Association (also known as the Category 6astandard), in the connection region where the blades of a modular plugmate with the contacts of the modular jack (referred to herein as the“plug-jack mating region”), the eight conductors 1-8 must be aligned ina row, with the eight conductors 1-8 arranged as four differential pairsspecified as depicted in FIG. 2. As is apparent from FIG. 2, thisarrangement of the eight conductors 1-8 will result in unequal couplingbetween the differential pairs, and hence both NEXT and FEXT isintroduced in each connector in industry standardized communicationssystems.

As the operating frequencies of communications systems has increased,crosstalk in the plug and jack connectors has became a more significantproblem. To address this problem, communications jacks now routinelyinclude compensating crosstalk circuits that introduce compensatingcrosstalk that was used to cancel much of the “offending” crosstalk thatis introduced in the plug jack mating region as a result of theindustry-standardized connector configurations. Initially, so-called“single-stage” crosstalk compensation circuits were developed that couldcancel the “offending” crosstalk that is generated in a plug jackconnector because a first conductor of a first differential pair couplesmore heavily with a first of the two conductors of a second differentialpair than does the second conductor of the first differential pair.Typically, these single-stage crosstalk compensation circuits wereimplemented by configuring the jack so that the second conductor of thefirst differential pair would couple with the first of the twoconductors of the second differential pair later in the jack to providea “compensating” crosstalk signal. As the first and second conductors ofthe differential pair carry equal magnitude, but opposite phase signals,so long as the magnitude of the “compensating” crosstalk signal that isinduced in such a fashion is equal to the magnitude of the “offending”crosstalk signal, then the compensating crosstalk signal that isintroduced later in the jack may substantially cancel out the offendingcrosstalk signal.

While the above-described “single-stage” crosstalk compensation circuitswere generally effective at cancelling out most of the crosstalk for lowfrequency signals (e.g., below 100 MHz), as the industry moved to higherfrequency signals the phase change between the offending crosstalksignal and the compensating crosstalk signal became more significantsuch that it became difficult to achieve sufficient crosstalkcompensation. Consequently, the use of “multi-stage” crosstalkcompensation schemes became common. Such crosstalk schemes are describedin U.S. Pat. No. 5,997,358 to Adriaenssens et al., the entire content ofwhich is hereby incorporated herein by reference as if set forth fullyherein.

Work is now ongoing in the industry to develop a Category 8 standardthat will specify parameters for higher data rate communications plugs,jacks and cable segments that may operate at higher frequencies. Forexample, the above-referenced ANSI/TIA-568-C.2 Category 6a standardprovides for communications at frequencies up to 500 MHz. In contrast,it is anticipated that the Category 8 standard may call forcommunications at frequencies up to, for example, 2 GHz. Moreover, it isanticipated that Category 8 connectors (e.g., plug and jacks) may berequired to exhibit full backwards compatibility so that they may beused with conventional Category 6 or 6a connectors while meeting thecomponent and channel performance requirements set forth in the Category6 and 6a standards. Special challenges may be involved in providingcommunications connectors that can meet the Category 8 performancestandards over the full anticipated Category 8 frequency range whilealso providing full backwards compatibility.

SUMMARY

Pursuant to embodiments of the present invention, patch cords areprovided that include a communications cable that has at least a firstconductor, a second conductor, a third conductor and a fourth conductor.A plug is attached to a first end of the communications cable. This plugincludes a housing that receives the communications cable and firstthrough fourth plug contacts that are at least partially within thehousing. First through fourth conductive paths connect respective onesof the conductors to respective ones of the plug contacts. The first andsecond conductors, the first and second conductive paths, and the firstand second plug contacts form a first differential transmission linethrough the plug, and the third and fourth conductors, the third andfourth conductive paths, and the third and fourth plug contacts form asecond differential transmission line through the plug. The firstdifferential transmission line includes at least a first segment havinga first impedance, a second segment having a second impedance, and athird segment having a third impedance, where the second segment isbetween the first and third segments, and where the first impedance isdifferent than the second impedance and the second impedance isdifferent than the third impedance. The first, second and thirdimpedances and the electrical lengths of the first, second and thirdsections are selected to provide a desired return loss spectrum for thefirst differential transmission line.

In some embodiments, the local maximum in the return loss spectra of thefirst differential transmission line may be positioned within the returnloss spectra in order to extend the operating frequency range of thepatch cord over which a minimum return loss margin may be maintained. INsome embodiments, the local maximum in the return loss spectra of thefirst differential transmission line may be within a pre-selectedfrequency range of an operating frequency range of the patch cord. ormay be at a frequency that is outside the operating frequency range ofthe patch cord but that is no more that 50% higher than the highestfrequency in the operating frequency range of the patch cord.

In some embodiments, the first through fourth conductive paths eachtraverse a printed circuit board, and the first segment of the firstdifferential transmission line connects to the second segment of thefirst differential transmission line on the printed circuit board. Thesecond segment of the first differential transmission line may connectto the third segment of the first differential transmission line on theprinted circuit board. A first pair of conductive trace sections on theprinted circuit board that form the first segment of the firstdifferential transmission line may have widths and/or thicknesses thatare different than the corresponding widths and/or thicknesses of asecond pair of conductive trace sections on the printed circuit boardthat form the second segment of the first differential transmissionline.

In some embodiments, the local maximum in the return loss spectra of thefirst differential transmission line is between a frequency of 1 GHz anda frequency of 5 GHz. The plug may be an RJ-45 plug. The impedance ofthe first segment of the first differential transmission line may differfrom the impedance of the second segment of the first differentialtransmission line by at least 20 percent, and the impedance of thesecond segment of the first differential transmission line may differfrom the impedance of the third segment of the first differentialtransmission line by at least 20 percent.

Pursuant to further embodiments of the present invention, communicationsplugs are provided that include a housing that receives the conductorsof the communication cable. A printed circuit board is mounted at leastpartially within the housing. A plurality of plug contacts are on theprinted circuit board, and the printed circuit board includes aplurality of conductive paths that electrically connect respective onesof the conductors to respective ones of the plug contacts. First andsecond of the conductive paths are arranged as a first differential pairof conductive paths that comprise a portion of a first differentialtransmission line through the communications plug, where the firstdifferential transmission line includes a first transition region wherethe impedance of the first differential transmission line changes by atleast 20%.

In some embodiments, the first differential transmission line mayinclude a second transition region where the impedance of the firstdifferential transmission line changes by at least 20%. The magnitude ofthe impedance change at the first transition region and the magnitude ofthe impedance change at the second transition region may be selected toimprove the return loss of the first differential transmission line inthe frequency range of 1 GHz to 2 GHz. The impedance change at the firsttransition region may be created at least in part by varying the widths,thicknesses and/or the spacing of the conductors used to form the firstdifferential transmission line adjacent the first transition region, byvarying the distance(s) of the conductors from adjacent image planes, ifany, by varying the dielectric constants of the surrounding materials,by adding one or more capacitances to one or both members of thedifferential transmission line, and/or by adding one or more inductancesin series with one or both members of the differential transmissionline. The return loss spectra of the first transmission line may includea local maximum at a frequency above 500 MHz.

Pursuant to further embodiments of the present invention, methods ofimproving the return loss on a differential transmission line through acommunications connector are provided in which the differentialtransmission line is divided into at least a first segment, a secondsegment and a third segment, wherein a first impedance mismatch betweenthe impedances of the first and second segments differs by at least 20%,and a second impedance mismatch between the impedances of the second andthird segments differs by at least 20%.

Pursuant to still further embodiments of the present invention,communications connectors are provided that include a printed circuitboard having a plurality of input terminals, a plurality of outputterminals, and a plurality of conductive paths that connect each inputterminal to a respective output terminal, wherein the conductive pathsare arranged as a plurality of differential transmission lines; and asolenoid inductor implemented along at least one of the conductivepaths.

In some embodiments, the solenoid inductor may comprise a first tracesegment on a first layer of the printed circuit board, a second tracesegment on a second layer of the printed circuit board, a third tracesegment on the first layer of the printed circuit board adjacent thefirst trace segment, a first conductive via connecting the first tracesegment to the second trace segment and a second conductive viaconnecting the second trace segment to the third trace segment.

In some embodiments, the solenoid inductor may comprise a first solenoidinductor that is implemented on a first of the conductive paths, thecommunications connector further comprising a second solenoid inductorimplemented along a second of the conductive paths, the first and secondconductive paths together forming a differential transmission line. Insome embodiments, the communications connector may be an RJ-45 plug oran RJ-45 plug. In some embodiments, the second trace segment mayvertically overlap at least one of the first trace segment or the thirdtrace segment. In some embodiments, the first trace segment may extendin parallel to the third trace segment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic diagram illustrating the use ofconventional communications plugs and jacks to interconnect a computerwith network equipment.

FIG. 2 is a schematic diagram illustrating the modular jack contactwiring assignments for a conventional 8-position communications jack(TIA 568B) as viewed from the front opening of the jack.

FIG. 3 is a circuit diagram of a model of a conventional differentialtransmission line through a communications plug.

FIG. 4A is a graph illustrating the return loss of a first plug thatdoes not use the return loss improvement techniques according toembodiments of the present invention and of a second plug that uses thereturn loss techniques according to embodiments of the presentinvention.

FIG. 4B is a graph illustrating the insertion loss of a first plug thatdoes not use the return loss improvement techniques according toembodiments of the present invention and of a second plug that uses thereturn loss techniques according to embodiments of the presentinvention.

FIG. 5 is a circuit diagram of a model of a differential transmissionline through a communications plug that includes impedance imbalancesthat improve return loss performance over a selected frequency band.

FIG. 6 is a graph that illustrates the currents into a load capacitor ofthe transmission line of FIG. 5 as a function of frequency.

FIG. 7 is a perspective view of a patch cord according to certainembodiments of the present invention.

FIG. 8 is a top-rear perspective view of a plug that is included on thepatch cord of FIG. 7.

FIG. 9 is a bottom-rear perspective view of the plug of FIG. 8.

FIG. 10 is a side view of the plug of FIG. 8.

FIGS. 11-14 are various perspective views of a printed circuit board ofthe plug of FIGS. 7-10.

FIG. 15 is a graph illustrating the simulated return loss performance ofthe communications plug of FIGS. 8-14.

FIG. 16 is a circuit diagram of a model of a differential transmissionline through a communications plug that illustrates how the impedanceimbalances may alternatively be implemented using discrete reactivecomponents.

FIG. 17 is a graph illustrating the close match in the return lossperformance of the transmission lines of FIGS. 5 and 16.

FIG. 18 is a flow chart illustrating operations for designing atransmission line through a communications connector according toembodiments of the present invention.

FIG. 19 is a circuit diagram of a model of a conventional Cat 6A testplug.

FIG. 20 is a graph illustrating the simulated return loss performance ofthe transmission lines of the communications plug of FIG. 19.

FIG. 21 is a circuit diagram of a model of the Cat 6A test plug of FIG.19 that further includes discrete elements that provide return loss andinsertion loss enhancements.

FIG. 22 is a graph illustrating the simulated return loss performance ofthe transmission lines of the communications plug of FIG. 21.

FIG. 23 is a graph illustrating the simulated return loss performance ofthe transmission lines of a model of the Cat 6A test plug of FIG. 19that includes both discrete elements and transmission line impedanceimbalances that together provide return loss and insertion lossenhancements.

FIG. 24 is a top view of a pair of printed circuit board based solenoidinductors that may be used in communications connectors according toembodiments of the present invention.

FIG. 25 is an oblique view of the printed circuit board based solenoidinductors of FIG. 24.

DETAILED DESCRIPTION

Pursuant to embodiments of the present invention, communicationsconnectors such as communications plugs are provided that may meetcrosstalk as well as return loss and insertion loss performancerequirements.

As noted above, in communications connectors that include multipledifferential pairs, crosstalk is an important performance parameter thatimpacts the throughput (data rate) that the connector can accommodate.Another important parameter in communications connectors is the returnloss that is experienced along each differential pair (i.e.,differential transmission line) through the connector. The return lossof a transmission line is a measure of how well the transmission line isimpedance matched with a terminating device or with loads that areinserted along the transmission line. In particular, the return loss isa measure of the signal power that is lost due to signal reflectionsthat may occur at discontinuities (impedance mismatches) in thetransmission line. Return loss is typically expressed as a ratio indecibels (dB) as follows:

${{RL}\mspace{11mu} ({dB})} = {10\mspace{11mu} \log_{10}\frac{P_{i}}{P_{r}}}$

where RL(dB) is the return loss in dB, P_(i) is the incident power andP_(r) is the reflected power. High return loss values indicate a goodimpedance match (i.e., little signal loss due to reflection), whichresults in lower insertion loss values, which is desirable. The industrystandards will typically specify minimum return loss requirements forthe transmission lines within individual connectors, within matedconnectors (i.e., across a mated plug and jack) and/or for an entirecommunications channel (i.e., for one or more differential transmissionlines that extend from computer 11 to server 20 in FIG. 1 across variousconnectors and cable segments). As return loss typically decreases withincreasing frequency (i.e., the return loss performance gets worse withincreasing frequency), the industry standards typically specify minimumreturn loss values that must be met as a function of frequency for thespecified components and/or channels.

In communications systems that include multiple differentialtransmission lines, it is commonplace for inductive and capacitivecouplings to exist between various of the transmission lines. Forexample, as discussed above, offending crosstalk and compensatingcrosstalk circuits are routinely provided in plug and jack connectors.Unfortunately, these inductive and capacitive couplings appear as loadsalong the transmission line that can degrade the return loss of thetransmission line.

By way of example, the Category 6 and 6a standards require thatstandards-compliant communications plugs introduce predetermined amountsof “offending” crosstalk between the four differential pairs thereof. Ifthe Category 8 standard requires backwards compatibility with theCategory 6 and 6a standards, then it will be necessary for Category 8standards-compliant plugs to inject the industry-standardized amounts ofoffending crosstalk between the four differential transmission lines aswell. However, this “offending” crosstalk will appear as loads on eachof the differential transmission lines through the plug. Moreover, inorder to comply with the crosstalk requirements set forth in theCategory 6a standard, it will likely also be necessary to includecrosstalk compensation circuits in the Category 8 communications jacksthat substantially cancel the offending crosstalk that is generated inthe plug. This compensating crosstalk thus will also appear as loads onthe four differential transmission lines that pass through the matedplug jack connectors. It is anticipated that it may be difficult tomaintain acceptable return loss performance at higher frequencies (e.g.,frequencies above 1 GHz) in a communications plug (or mated plug jackcombination) while also injecting the necessary amount of offendingcrosstalk between the differential pairs and compensating for the samein the communications jack.

Pursuant to embodiments of the present invention, communicationsconnectors are provided that have signal paths that include one moreelectrical circuits that create resonances at one or more frequencies.These resonances (including the frequency range over which theresonances occur) may be tuned to enhance the return loss and/or theinsertion loss of the signal path within a desired range of frequencieswithout unacceptably degrading other characteristics of the signal path.The resonant frequencies are not necessarily within the range offrequencies within which return loss and/or insertion loss is enhanced.In fact, in some embodiments the resonance may be so wide that it doesnot create a peak or local maximum in either the return loss orinsertion loss spectra of the signal path. The electrical circuits thatgenerate the resonances can be implemented using, for example, discretecapacitors and inductors and/or using a transmission line containing atleast two impedance discontinuities of particular magnitudes separatedby a particular electrical length.

In some embodiments, communications connectors are provided that have atleast one transmission line that includes pre-selected impedancemismatches that are used to improve the return loss and/or the insertionloss of the transmission line over a desired frequency range. Inparticular, the transmission line may include at least two locationswhere impedance mismatches are provided that create resonances. Themagnitude of these impedance mismatches and the locations of theimpedance mismatches (which determines the time delay between theimpedance mismatches and other elements along the transmission line) maybe selected to improve the return loss and/or the insertion loss of thetransmission line over a selected frequency range such as, for example,from about 1 GHz to about 2 GHz. These designed impedance mismatches andassociated delay(s) may create resonances or other effects, includinglarge amounts of signal reflections, at frequency ranges outside of thefrequency range of interest. The resonances may be tuned by adjustingthe magnitudes of the impedance mismatches and the electrical delaysbetween the impedance mismatches. In some cases, the resonances maycreate a local maximum in the return loss spectra and/or a local minimumin the insertion loss spectra for the transmission line. These localmaxima and minima may (but need not be) within or just outside thefrequency range for which the transmission line is designed to operateand may be used to effectively extend the frequency range (to higherfrequencies) over which the transmission line may provide suitablereturn loss performance.

In some embodiments, the connector may be a communications plug. Theplug may be designed to have at least two impedance mismatches along atleast one of the differential transmission lines through the plug. Theseimpedance mismatches can be implemented in a wide variety of ways. Forexample, in plugs in which at least part of the transmission lines runacross (and perhaps through) a printed circuit board, one or moreimpedance mismatches may be created by, for example, (1) changing thewidth, thickness or spacing of the conductive traces/elements on theprinted circuit board that form the transmission line segments, (2)varying the distance(s) of the conductors from adjacent image planes, ifany, and/or (3) by varying the dielectric constants of the surroundingmaterials. Impedance mismatches may also or alternatively be implementedat transitions within the plug such as the transition from conductivewires of a communications cable to printed circuit board conductivetraces or from printed circuit board conductive traces to the blades ofthe plug. By carefully selecting the magnitude of these impedancemismatches and the delays between the mismatches it is possible tosignificantly improve the return loss performance of the transmissionline over a selected frequency range.

As noted above, in other embodiments, the time delayed impedancemismatches along the transmission line(s) of the communicationsconnector may be replaced by discrete reactive elements such ascapacitors and/or inductors. For example, on a balanced (differential)transmission line discrete reactive elements may be provided along thetransmission line that resemble and/or function as a “box-section”filter (i.e., two shunt capacitors between the conductors of thetransmission line with a series inductor along each conductor betweenthe two capacitors). For connectors that use single-ended transmissionlines, the discrete reactive elements may be arranged as a pi-sectionfilter (i.e., two shunt capacitors with a series inductor therebetween).In still other embodiments, a combination of the time-delayed impedancemismatches and discrete reactive elements may be used to generate theresonances that are tuned to provide improved return loss and insertionloss performance.

In some embodiments, the connectors may comprise Category 8 standardplugs that include coupling components such as capacitors or inductivecoupling sections that are used to increase the crosstalk between thedifferential transmission lines through the plug in order to comply withthe offending crosstalk levels specified in the Category 6a standard. Asnoted above, such coupling components may be provided to ensure that theplug is backwards compatible with the Category 6a standard. In someembodiments, these crosstalk coupling components may be incorporatedinto the electrical circuit that is provided to improve return lossand/or insertion loss performance. In other embodiments, the connectorsmay comprise Category 6a plugs. It will also be appreciated thatembodiments of the present invention are not limited to Ethernetconnectors or to connectors with differential transmission lines.Additionally, while embodiments of the present invention are primarilydiscussed herein with respect to communications plugs, it will beappreciated that crosstalk compensation is routinely introduced in jacksand that the techniques that are described herein may also be used injacks to improve the return loss and/or insertion loss performancethereof.

Embodiments of the present invention will now be discussed in greaterdetail with reference to the drawings.

FIG. 3 is a simplified circuit model of a single-ended (i.e.,non-differential) 50 ohm transmission line 100 that illustrates howcrosstalk circuits (i.e., offending crosstalk or compensating crosstalkcomponents, whether or not intentionally introduced) can create a loadalong a transmission line. FIG. 4A is a graph illustrating the modeledreturn loss of the transmission line 100 of FIG. 3 that illustrates howthe transmission line 100 may fail to meet the proposed return lossparameter for the Category 8 standard at high frequencies. FIG. 4A alsoincludes a graph illustrating the modeled return loss of a transmissionline according to embodiments of the present invention that illustratesthe degree to which return loss performance may readily be improved.This graph in FIG. 4A will be discussed below in conjunction with FIG.5. FIG. 4B is a graph illustrating the modeled insertion loss of thetransmission line 100 of FIG. 3 as compared to the modeled insertionloss of a transmission line according to embodiments of the presentinvention.

As shown in FIG. 3, the transmission line 100 may be modeled as a signalsource with internal series termination 110, a transmission line segment120, a capacitive load 130 and an end termination 115. Each of theterminations 110, 115 are 50 ohm terminations, as is the transmissionline segment 12Q. The transmission line segment 120 has a length thatresults in a delay of 0.083 nanoseconds (i.e., an RE signal willtraverse each transmission line segment in 0.083 nanoseconds). Thecapacitive load 130 is assumed to be a shunt 1.2 pF capacitance, whichis representative of the type of capacitive load that results from theoffending crosstalk requirements in the Category 6a standard. While thecrosstalk load 130 is modeled as solely comprising capacitive crosstalk,it will be appreciated that the pair-to-pair crosstalk in communicationsconnectors will typically include both a capacitive component and aninductive component. The inductive component of the crosstalk maylikewise appear as a load on the transmission line and have similareffects on return loss performance. The effect of the crosstalk ismodeled in FIG. 3 as a purely capacitive effect in order to simplify theexample and associated modeling.

Referring now to FIG. 4A, one proposed return loss specification for atransmission line through a Category 8 plug is shown as curve 150. Asshown in FIG. 4A, a return loss of at least 33 dB is specified at 100MHz and this minimum return loss requirement drops to about 10 dB at 1.5GHz, where it remains constant. As is also shown in FIG. 4, the returnloss performance of the transmission line 100 of FIG. 3 (curve 160) hasa margin of about 1 dB for frequencies below 1.5 GHz, but at frequenciesabove 1.7 GHz the return loss of the transmission line 100 falls belowthe proposed standard (curve 150). This degradation in return lossperformance at higher frequencies may be attributable to the loadimparted by the capacitor 130 on the transmission line 100. As thecapacitive load 130 may be necessary to comply with industry standardsrequirements (e.g., interface specifications, crosstalk specifications),it may be difficult to meet return loss performance standards at higherfrequencies, as shown in FIG. 4A.

FIG. 5 illustrates how the techniques according to embodiments of thepresent invention may be used to improve the return loss of thetransmission line 100 of FIG. 3 so that it meets the proposed Category 8return loss standard with significant margin.

As shown in FIG. 5, the transmission line 100 of FIG. 3 may be replacedwith a transmission line 100′. The transmission line 100′ includes thesame signal source with internal series termination 110, the same endtermination 115 and the same capacitive load 130 that are included inthe transmission line 100 of FIG. 3. However, the 50 ohm transmissionline segment 120 of the transmission line 100 of FIG. 3 is replaced witha pair of transmission line segments 120′ and 125′. The impedances ofthe transmission line segments 120′ and 125′ and the lengths of thesetransmission line segments are selected to create impedance mismatchesthat impact the return loss of transmission line 100′ in a desiredmanner. In this example, the transmission line segment 120′ has animpedance of 31 ohms, and a delay of 0.043 nanoseconds, and thetransmission line segment 125′ has an impedance of 96 ohms, and a delayof 0.04 nanoseconds. Thus, a first impedance mismatch 121 is generatedat the interface between the termination 110 and the transmission linesegment 120′ (where a 50 ohm element and a 31 ohm element interface), asecond impedance mismatch 122 is generated at the interface between thetransmission line segment 120′ and the transmission line segment 125′(where a 31 ohm element and a 96 ohm element interface), and a thirdimpedance mismatch 123 is generated at the interface between thetransmission line segment 125′ and the end termination 115 (where a 96ohm element and a 50 ohm element interface).

Referring again to FIG. 4A, the curve 170 illustrates the modeled returnloss performance of the transmission line 100′. As shown in FIG. 4A,improved return loss is seen at all frequencies below 2.7 GHz, with thereturn loss performance dropping off more slowly at frequencies between100 MHz and 1 GHz as compared to the return loss performance oftransmission line 100 (curve 160), and then the return loss performanceactually improving for frequencies between about 1 GHz and 1.8 GHzbefore trending downward again. While Applicant does not intend to belimited to any particular theory of operation, it is believed that theimpedance mismatches at each of the interfaces 121-123 in thetransmission line 100′ may act like a low pass filter. This may, in somecases, generate a local peak or “maximum” in the return loss spectra(i.e., a plot of return loss as a function of frequency) as illustratedat reference numeral 175 in FIG. 4A. It will also be appreciated thatwhile the simplest designs may have low-pass filter characteristics,other designs that more resemble high-pass filters, band-stop filters orband-pass filters may also be used. While the return loss may, in somecases, fall off rapidly on the high frequency side of the local peak175, the provision of a local peak 175 in the return loss spectra mayallow for significantly improved return loss performance within afrequency range of interest. For example, as shown in FIG. 4A, bydesigning the transmission line 100′ so that the local peak 175 ispositioned, for example, at a frequency that is in the upper portion ofthe frequency range of operation (here, below 2 GHz), excellent returnloss performance may be achieved in many cases across the entireoperating frequency band.

An ideal, unloaded signal path (not shown in the figures) maytheoretically have essentially infinite return loss (i.e., no reflectionor signal degradation whatsoever). As shown in FIG. 4A, the loadimparted on the transmission line 100 by the load capacitor 130 degradesthe return loss at all frequencies, and this degradation reachesunacceptable levels (for this particular standard) at frequencies justbelow 2 GHz. As is also shown in FIG. 4A, the resonance created by theimpedance mismatches (see curve 170) can be tuned to enhance return lossperformance, especially in the frequency range of about 1 GHz to about 2GHz, where the crosstalk compensation circuits in a mated plug jackconnector would otherwise significantly degrade return loss performance.By designing the transmission line so that the resonances at leastmostly occur at frequencies that are outside the frequency range ofinterest (i.e., the frequency range that the connector is designed tosupport communications over), these resonances do not unduly impact theperformance of the transmission line. Moreover, as noted above, byvarying the magnitude of the impedance mismatches, the number ofimpedance mismatches and/or the delay between the impedance mismatchesit is possible in some embodiments to create local peaks in the returnloss spectra within the frequency band of interest. The inclusion ofsuch local peaks in the return loss spectra may be particularlyeffective in providing a transmission loss that has significantlyimproved return loss performance.

In some embodiments, the local maximum in the return loss spectra of thedifferential transmission line may be positioned within the return lossspectra in order to extend the operating frequency range of the patchcord over which a minimum return loss margin may be maintained. Forexample, the peak 175 is positioned near the upper end of the desiredoperating range (i.e., near 2 GHz) in order to improve the return lossmargin at higher frequencies. In some embodiments, the peak may bepositioned in the upper half of the desired frequency operation range(i.e., between 1 GHz and 2 GHz in the example of FIG. 4A). In otherembodiments, the peak may be positioned at a somewhat frequency abovethe highest frequency in the desired frequency operation range. Forexample, the peak may be positioned at a frequency that is outside thedesired frequency operating range but that is no more that 50% higherthan the highest frequency in the desired frequency operation range(i.e., between 2 GHz and 3 GHz in the example of FIG. 4A).

Referring next to FIG. 4B, the curve 180 illustrates the modeledinsertion loss performance of the transmission line 100 and the curve185 illustrates the modeled insertion loss performance of thetransmission line 100′. As shown in FIG. 4B, significant improvements ininsertion loss may also be achieved by the techniques according toembodiments of the present invention. Similar to the return loss spectraof FIG. 4A, the techniques according to embodiments of the presentinvention may provide a local minimum in the insertion loss (identifiedby reference numeral 187 in FIG. 4B). In some embodiments, this localminimum may be positioned, for example, at a frequency that is in theupper portion of the frequency range of operation (here, below 2 GHz) inorder to provide excellent insertion loss performance across the entireoperating frequency band. As shown in FIG. 4B, insertion loss may beimproved by 0.5 dB or more in certain frequency ranges.

FIG. 6 illustrates the modeled current flow in the transmission line100′ of FIG. 5. In particular, a current probe was included in thecircuit model of FIG. 5 that measured the current flow, as a function offrequency, into the load capacitor 130. In FIG. 6, curve 190 is thesimulated current flow into the load capacitor 130 in the transmissionline 100 of FIG. 3, while curve 195 is the simulated current flow intothe load capacitor 130 in the transmission line 100′ of FIG. 5.

As shown in FIG. 6, no resonance is found in the current flow into theload capacitor 130 of the transmission line 100 (see curve 190).However, a resonance appears in the current flow into the load capacitor130 in the transmission line 100′ of FIG. 5 (see curve 195). Thisresonance is attributable to the impedance mismatch created by the loadimparted on the transmission line 100′ by the capacitor 130 togetherwith the electrical length (time delay) of the intentionally addedimpedance mismatches and their associated time delays along thetransmission line 100′. As shown in FIG. 6, the current into the loadcapacitor 130 may vary widely with frequency. In some embodiments, thetransmission line 100′ may be designed so that these resonances occur atfrequencies that are outside the frequency range of interest (herefrequencies above 2 GHz). This may facilitate ensuring that theresonances do not negatively affect the performance of the transmissionline 100′. Moreover, by varying the magnitude of the impedancemismatches and/or the delay between these impedance mismatches it ispossible to create one or more local peaks in the return loss spectra(see peak 175 in FIG. 4A) and/or one or more local minimums in theinsertion loss spectra (see valley 187 in FIG. 4B) within or near thefrequency band of interest, thereby providing a transmission line thatmay exhibit significantly improved return loss and/or insertion lossperformance.

The impedance mismatches can be created in a wide variety of ways. Asknown to those of skill in the art, the impedance of a differentialtransmission line will be a function of, among other things, the shapeof the conductors, the size of the conductors, the material used to formthe conductors, the spacing between the conductors, the dielectricconstants of the materials surrounding and between the conductors, etc.Thus, the above-described impedance mismatches may be implemented bychanging any of these parameters. Trial and error or more deliberatemodeling techniques may be used to identify impedance mismatches thatprovide a desired improvement in return loss performance over a selectedfrequency band.

For example, computer programs may be used to model the return loss (orinsertion loss) of a transmission line through a plug that has one prmore loads imparted thereon. The modeling may take into account not onlythe loads injected on the transmission line by crosstalk circuits andother closely-spaced transmission lines, but may also take into accountimpedance mismatches that will occur in transitions along thetransmission line where, for example, a conductive wire connects to aprinted circuit board trace, where insulation is removed from aconductive wire or where a conductive wire mates with a plug blade. Oneor more of the transmission line segments that extend between the loadsmay then be changed in the model to have a different impedance and/or adifferent length. A trial and error process may be used where thelengths and/or the size of the impedance mismatches are varied. Thechanges in return loss tend to be predictable as this trial and errorprocess is carried out, so a circuit designer can readily identifyimpedance mismatches and transmission line segment lengths that willprovide improved performance over a frequency range of interest. Inother embodiments, this process could be automated using, for example,commercially available programs that find local maximum in an outputparameter when a variety of input parameters are varied.

In some embodiments, the amount of the impedance mismatch may be atleast about 20%. In other embodiments, the amount of the impedancemismatch may be at least about 30%. In still other embodiments, theamount of the impedance mismatch may be at least about 40%. Herein,where references are made to changes in impedance or to impedancemismatches that are specified using a percentage, the percentage iscalculated by dividing the difference between the two impedances by thelarger of the two impedances. For example, if a first transmission linesegment having an impedance of 80 ohms connects to a second transmissionline segment that has an impedance of 100 ohms, the change in impedancebetween these two transmission line segments or, equivalently, thedegree of impedance mismatch, would be (100 ohms-80 ohms)/100 ohms=20%.

FIGS. 7-14 illustrate a patch cord 200 and various components thereofaccording to certain embodiments of the present invention. Inparticular, FIG. 7 is a perspective view of the patch cord 200. FIG. 8is a top-rear perspective view of a plug 216 that is included on thepatch cord 200 of FIG. 7. FIG. 9 is a bottom-rear perspective view ofthe plug 216. FIG. 10 is a side view of the plug 216. FIGS. 11-14 arevarious perspective views of a printed circuit board 250 of plug the 216of FIGS. 7-10 that illustrate how the conductors 201-208 of the patchcord 200 connect to the plug blades 241-248 that are mounted on theprinted circuit board 250.

As shown in FIG. 7, the patch cord 200 includes a cable 209 that haseight insulated conductors 201-208 enclosed in a jacket 210 (note thatthe conductors 201-208 are not individually numbered in FIG. 7, andconductors 204 and 205 are not visible in FIG. 7). The insulatedconductors 201-208 may be arranged as four twisted pairs of conductors211-214 (pair 211 is not visible in FIG. 7), with conductors 204 and 205twisted together to form twisted pair 211, conductors 201 and 202twisted together to form twisted pair 212, conductors 203 and 206twisted together to form twisted pair 213, and conductors 207 and 208twisted together to form twisted pair 214. A separator 215 such as atape separator or a cruciform separator may be provided that separatesone or more of the twisted pairs 211-214 from one or more of the othertwisted pairs 211-214. A first plug 216 is attached to a first end ofthe cable 209 and a second plug 218 is attached to the second end of thecable 209 to form the patch cord 200. Strain relief boots (not shown inFIG. 7) may be attached to each of the plugs 216, 218 which resist thetendency for a longitudinal force applied to the cable 209 to pull thecable 209 out of the plugs 216, 218.

FIGS. 8-10 are enlarged views that illustrate the first plug 216 of thepatch cord 200. In order to simplify the drawing, a rear cap of the plughousing, various wire grooming and wire retention mechanisms and thestrain relief boot are not shown in FIGS. 8-10. As shown in FIGS. 8-10,the communications plug 216 includes a housing 220 that has a bi-leveltop face 222, a bottom face 224, a front face 226, and a rear opening228 that receives a rear cap (not shown). A plug latch 232 extends fromthe bottom face 224. The top and front faces 222, 226 of the housing 220include a plurality of longitudinally extending slots 234. Thecommunications cable 209 (see FIG. 7) is received through the rearopening 228. A rear cap (not shown) that includes a cable aperture locksinto place over the rear opening 228 of housing 220 after thecommunications cable 209 has been inserted therein.

As is also shown in FIGS. 8-10, the communications plug 216 furtherincludes a printed circuit board 250 which is disposed within thehousing 220, and a plurality of plug blades 241-248 are mounted at theforward edge of the printed circuit board 250 so that the blades 241-248can be accessed through the slots 234 in the top face 222 and front face226 of the housing 220. The housing 220 may be made of a suitableinsulative plastic material that meets applicable standards with respectto, for example, electrical breakdown resistance and flammability suchas, for example, polycarbonate, ABS, ABS/polycarbonate blend or otherdielectric molded materials. Any conventional housing 220 may be usedthat is configured to hold the printed circuit board 250, and hence thehousing 220 is not described in further detail herein.

FIGS. 11 and 12 are enlarged perspective top and bottom views,respectively, of the printed circuit board 250 and the plug blades241-248 that illustrate these structures in greater detail and that showhow the insulated conductors 201-208 of communications cable 209 may beelectrically connected to the respective plug blades 241-248 through theprinted circuit board 250. FIGS. 13 and 14 are enlarged perspective topand bottom views, respectively, of the top and bottom surfaces of theprinted circuit board 250 and the plug blades 241-248. In FIGS. 13 and14, the dielectric portion of the printed circuit board 250 is omittedin order to better illustrate certain features of the printed circuitboard 250. In FIG. 13, only the base portions of the plug blades 241-248are shown in order to illustrate how the base portion of each plug blade241-248 is received within a respective one of a plurality ofmetal-plated vias 231-238 in the printed circuit board 250 in order tomount the plug blades 241-248 on the printed circuit board 250.

The printed circuit board 250 may comprise, for example, a conventionalprinted circuit board, a specialized printed circuit board (e.g., aflexible printed circuit board) or any other appropriate type of wiringboard. In the embodiment of the present invention depicted in FIGS.8-14, the printed circuit board 250 comprises a conventional multi-layerprinted circuit board.

As shown in the figures, the printed circuit board 250 includes fourplated pads 251, 252, 254, 255 on a top surface thereof and anadditional four plated pads 253, 256-258 on a bottom surface thereof.The insulation is removed from an end portion of each of the conductors201-208 of the communications cable 209, and the metal (e.g., copper)core of each conductor 201-208 may be soldered, welded or otherwiseattached to a respective one of the plated pads 251-258. By terminatingeach of the conductors 201-208 directly onto the plated pads 251-258without the use of any insulation displacement contacts, insulationpiercing contacts or other wire connection contacts, the size of theplug 216 may be reduced. It will also be appreciated that othertechniques may be used for terminating the conductors 201-208 to theprinted circuit board 250. For example, in other embodiments, aplurality of insulation piercing contacts or insulation displacementcontacts may be mounted on the printed circuit board 250 that are usedto electrically connect each conductor 201-208 to a respective trace onthe printed circuit board 250. It will be appreciated that in otherembodiments all of the conductors 201-208 may be mounted exclusively onthe bottom surface of the printed circuit board 250 or exclusively onthe top surface of the printed circuit board 250.

As is best shown in FIGS. 11-13, the conductors 201-208 may bemaintained in pairs within the plug 216. A cruciform separator 230 maybe included in the rear portion of the housing 220 that separates eachpair 211-214 from the other pairs 211-214 in the cable 209 to reducecrosstalk in the plug 216. The conductors 201-208 of each pair 211-214may be maintained as a twisted pair all of the way from the rear opening228 of plug 216 up to the printed circuit board 250.

The plug blades 241-248 are configured to make mechanical and electricalcontact with respective contacts, such as, for example, spring jackwirecontacts, of a mating communications jack. In particular, as shown bestin FIGS. 11-12 and 14, each of the eight plug blades 241-248 is mountedat the front portion of the printed circuit board 250. The plug blades241-248 may be substantially transversely aligned in side-by-siderelationship. Each of the plug blades 241-248 includes a first sectionthat extends forwardly along a top surface of the printed circuit board250, a transition section that curves through an angle of approximatelyninety degrees and a second section that extends downwardly from thefirst section along a portion of the front edge of the printed circuitboard 250. The transition section may include a curved outer radius thatcomplies with the specification set forth in, for example, IEC 60603-7-4for industry standards compliant plug blades.

Each of the plug blades 241-248 may be fabricated separately from theprinted circuit board 250. In the depicted embodiment, each of the plugblades 241-248 comprise, for example, an elongated metal strip having alength of approximately 140 mils, a width of approximately 20 mils and aheight (i.e., a thickness) of approximately 20 mils. Each plug blade241-248 also includes a base column 249 (see FIG. 13) that extends froma bottom surface of the plug blade. The printed circuit board 250includes eight metal-plated vias 231-238 that are arranged in two rowsalong the front edge thereof (only vias 231 and 238 are labeled in FIG.13 to simplify the drawing). The base column 249 of each plug blade241-248 is received within a respective one of the metal-plated vias231-238 where it may be press-fit, welded or soldered into place tomount the plug blades 241-248 on the printed circuit board 250.

The plug blades 241-248 may be mounted to the printed circuit board 250in other ways. For example, in other embodiments, elongated contact padsmay be provided on the top surface of the printed circuit board 250 andeach plug blade 241-248 may be welded or soldered to a respective one ofthese contact pads. It will be appreciated that many other attachmentmechanisms may be used.

Turning again to FIGS. 11-14 it can be seen that a plurality ofconductive paths 261-268 are provided on the top and bottom surfaces ofthe printed circuit board 250. Each of these conductive paths 261-268electrically connects one of the plated pads 251-258 to a respective oneof the metal-plated vias 231-238 so as to provide an electrical pathbetween each of the conductors 201-208 that are terminated onto theplated pads 251-258 and a respective one of the plug blades 241-248 thatare mounted in the metal-plated vias 231-238. Each conductive path261-268 may comprise, for example, one or more conductive traces thatare provided on one or more layers of the printed circuit board 250.When a conductive path 261-268 includes conductive traces that are onmultiple layers of the printed circuit board 250 (i.e., conductive paths264, 265 and 268 in the depicted embodiment), metal-plated ormetal-filled through holes (or other layer-transferring structures knownto those skilled in this art) may be provided that provide an electricalconnection between the conductive traces on different layers of theprinted circuit board 250.

A total of four differential transmission lines 271-274 are providedthrough the plug 216. The first differential transmission line 271includes the end portions of conductors 204 and 205, the plated pads 254and 255, the conductive traces 264 and 265, the plug blades 244 and 245,and the metal-plated vias 249 that are used to mount plug blades 244 and245. The second differential transmission line 272 includes the endportions of conductors 201 and 202, the plated pads 251 and 252, theconductive traces 261 and 262, the plug blades 241 and 242, and themetal-plated vias 249 that are used to mount plug blades 241 and 242.The third differential transmission line 273 includes the end portionsof conductors 203 and 206, the plated pads 253 and 256, the conductivetraces 263 and 266, the plug blades 243 and 246, and the metal-platedvias 249 that are used to mount plug blades 243 and 246. The fourthdifferential transmission line 274 includes the end portions ofconductors 207 and 208, the plated pads 257 and 258, the conductivetraces 267 and 268, the plug blades 247 and 248, and the metal-platedvias 249 that are used to mount plug blades 247 and 248.

As shown in FIGS. 11-14, the two conductive traces 261-268 that formeach of the differential transmission lines 271-274 on the printedcircuit board 250 are generally run together, side-by-side, on theprinted circuit board 250. Running the conductive traces 261-268 of eachdifferential transmission line 271-274 side-by-side may provide improvedimpedance matching so that each segment of a particular transmissionline may have a relatively constant impedance. This approach may make iteasier to model the performance of the transmission line and hence todesign a transmission line that meets pre-selected performance criteria.As shown best in FIGS. 13 and 14, the conductive traces 264, 265 thatare part of the transmission line 271 and the conductive traces 263, 266that are part of the transmission line 273 each traverse the printedcircuit board 250 at an angle with respect to the longitudinal axis ofthe printed circuit board 250. These angled conductive traces are merelyused for routing purposes to connect a pair of conductors (e.g.,conductors 204 and 205) that are located on one side of the longitudinalaxis of the printed circuit board 250 to the metal-plated vias 234, 235,which span the longitudinal axis of the printed circuit board 250.

A plurality of offending crosstalk circuits are also included on theprinted circuit board 250. In particular, a total of five offendingcrosstalk capacitors 281-285 are provided adjacent the plug blades241-248. Capacitor 281 injects offending crosstalk between blades 241and 242, capacitor 282 injects offending crosstalk between blades 242and 243, capacitor 283 injects offending crosstalk between blades 243and 244, capacitor 284 injects offending crosstalk between blades 245and 246, and capacitor 285 injects offending crosstalk between blades246 and 247. Additionally, an inductive coupling section 286 is includedbetween conductive traces 266 and 267 which injects offending inductivecrosstalk between transmission lines 273 and 274 in order to meet thespecified offending FEXT requirements in the Category 6a standard. Theoffending crosstalk circuits 281-286 may be provided, for example, toensure that the plug 216 meets all of the pair-to-pair offendingcrosstalk specifications required by an industry standards document suchas the aforementioned ANSI/TIA-568-C.2 standard. Unfortunately, theseoffending-crosstalk circuits 281-286 appear as loads along each of thetransmission lines 271-274 through the plug 216 that may make itdifficult for the plug 216 to meet target return loss performancespecifications, particularly at higher frequencies (e.g., frequenciesabove 500 MHz and even more so with respect to frequencies above 1 Ghzor above 1.5 GHz).

As discussed above, the communications connectors according toembodiments of the present invention may include two or more impedancemismatches along one or more of their transmission lines that improvethe return loss on each transmission line over a desired frequencyrange. These impedance mismatches may create resonances that may betuned to provide improved return loss and/or insertion loss performancefor the transmission line over selected frequency ranges. The impedancemismatches can be implemented, for example, by changing the width orthickness of the conductors that form the differential transmission linesegments or by changing the spacing of the conductors or the dielectricconstants of the insulating materials that are adjacent to theconductors. By carefully selecting the degree of these impedancemismatches and the delays between the mismatches it is possible tosignificantly improve the return loss performance of the transmissionline over a selected frequency range.

As is also shown in FIGS. 13 and 14, one or more reflection or “image”planes can be included in the plug 216. In the embodiment of FIGS. 7-14,two image planes 290, 292 are included in the plug 216, the first ofwhich is located just below a top surface of the printed circuit board250, and a second of which is located just above a bottom surface of theprinted circuit board 250. Each image plane 290, 292 may be implementedas a conductive layer on the printed circuit board 250. In someembodiments, the image planes 290, 292 may be grounded by, for example,connecting each image plane to a ground wire, drain wire or other groundreference so that each image plane 290, 292 will act as a ground plane.However, in other embodiments, the image planes 290, 292 may not beelectrically grounded (i.e., they are left floating electrically). Theimage planes 290, 292, whether or not they are electrically grounded,may act as shielding structures that reduce coupling between theportions of conductive traces 261-268 that are on the top side of theprinted circuit board 250 and the portions of conductive traces 261-268that are on the bottom side of the printed circuit board 250.

The image planes 290, 292 may also be used to control the impedance ofthe transmission lines 271-274. In particular, the impedance of eachdifferential transmission line is impacted by the distance of theconductors of the transmission line 271-274 from the image plane(s) 290,292. Thus, another way of creating impedance mismatches along one ormore of the transmission lines 271-274 in the plug 216 is to vary thedistance of the conductors that form the transmission line from one ofthe image planes 290, 292.

In the plug 216, the impedance mismatches are created by varying thewidths of the conductive traces 261-268, varying the spacings betweenthe conductive traces 261-268 and by varying the distances of theconductive traces from the image planes 290, 292 For example, conductivetrace 261 includes a first transition point 261-1 where the tracenarrows, a second transition point 261-2 where the trace narrowsfurther, and a third transition point 261-3 where the trace widens. Eachof these transition points 261-1 through 261-3 creates an impedancemismatch. Likewise, conductive trace 262 includes a first transitionpoint 262-1 where the trace narrows and a second transition point 262-2where the trace narrows further. These transition points create multipleimpedance mismatches along transmission line 272. Thus, multipleimpedance mismatches are created along transmission line 272.

In a similar fashion, transition points 263-1 and 263-2 are provided onconductive trace 263, and transition points 266-1 and 266-2 are providedon conductive trace 263. Additionally, near the front of the printedcircuit board 250 conductive traces 263 and 267 split apart in order tobe routed to their respective plug blades 243 and 246. This increase inthe spacing between conductive traces 263 and 266 create additionaltransition points 263-3 and 266-3. These transition points createmultiple impedance mismatches along transmission line 273.

Similarly, conductive trace 267 of transmission line 274 includes tworight angle turns adjacent the front edge of printed circuit board 250that vary the spacing between conductive path 267 and its counterpartconductive path 268. This variation in spacing creates additionalimpedance mismatches on conductive paths 267 and 268, respectively.Additional impedance mismatches are injected on transmission line 274via the transition points 267-1 and 268-1 that are provided on traces267 and 268. Thus, multiple impedance mismatches are also generatedalong transmission line 274.

Finally, conductive trace 264 includes a first transition point 264-1where the trace narrows, and conductive trace 265 likewise includes afirst transition point 265-1 where the trace narrows. These transitionpoints 264-1 through 265-1 create impedance mismatches. Anotherimpedance mismatch occurs in the conductive vias that are used totransition the conductive traces from the top side of printed circuitboard 250 to the bottom side of the printed circuit board 250. The wideseparation between conductive traces 264 and 265 near the front portionof printed circuit board 250 (i.e., on the bottom side of the board)also contributes to the impedance mismatch and further increasesinductive coupling between transmission lines 271 and 273 for purposesof meeting the crosstalk specifications for plug 216. Thus, multipleimpedance mismatches are also created along transmission line 271.

It can also be seen in FIGS. 13 and 14 that the image planes 290, 292 donot extent all the way forwardly to the front edge of the printedcircuit board 250. Another impedance mismatch is created where at thepoint where a conductive trace starts to extend beyond the front edge ofan image plan, such as at transition points 261-2 and 262-2 for theconductive traces of transmission line 272.

As discussed above, the number of impedance mismatches included on theconductors of a differential transmission line, the size of eachimpedance match, and the delays between impedance mismatches may beselected to enhance the return loss performance of the transmission linein a pre-selected frequency range. Trial and error approaches may beused to find a transmission line design that achieves desiredperformance levels, although computer programs may readily be developedthat may be used to better optimize return loss performance.

FIG. 15 is a graph illustrating the simulated return loss performance ofthe communications plug 216 of FIGS. 8-14. In FIG. 15, the solid curvesillustrate the simulated return loss on transmission lines 271-274 forsignals passing in the forward direction through plug 216 (i.e., fromthe cable attached to the plug 216 to a mating jack), while the curvesdrawn using dotted lines illustrate the simulated return loss ontransmission lines 271-274 for signals passing in the reverse directionthrough plug 216 (i.e., from a mating jack into the plug 216). Curve 294represents the minimum magnitude requirement for the Category 6astandard return loss (extrapolated above the Cat 6A requirement, whichends at 500 MHz). Curve 296 is the threshold above which the phase ofthe return loss need not be controlled. As is readily apparent, thesesimulations indicate that return loss margins of at least 6 dB areprovided on all four transmission lines 271-274 at frequencies above 500MHz.

The resonances that are used to improve return loss on the transmissionlines of connectors according to embodiments of the present inventionmay also be implemented using discrete elements. For example, FIG. 16illustrates an alternative circuit model for the transmission line ofFIG. 5. As shown in FIG. 16, the transmission line 100′ of FIG. 5 mayalternatively be implemented as a transmission line 300 which includes asignal source with internal series termination 310, a transmission linesegment 320, an end termination 315, and a pi filter 340 which comprisesa first 1.4 pF shunt capacitor 342, the capacitive load 330 (1.2 pF) anda series 4.2 nH inductor 344 therebetween Each of the terminations 310,315 are 50 ohm terminations, as is the transmission line segment 320.The transmission line segment 320 is modeled as having a length thatresults in a delay of 0.083 nanoseconds (i.e., an RF signal willtraverse each transmission line segment in 0.083 nanoseconds).

FIG. 17 illustrates the return loss of the transmission line 100′ ofFIG. 5 (curve 170) as compared to the return loss of the transmissionline 300 of FIG. 16 (curve 350). As is readily apparent, the two curvesare almost identical, which shows that the impedance mismatches create alow pass filtering effect that results in a local maximum in the returnloss spectra that may be used to improve the return loss of thetransmission line in a selected frequency range.

Pursuant to embodiments of the present invention, techniques aredisclosed for improving the return loss performance of communicationsconnectors, particularly at higher frequencies (e.g., frequencies above500 MHz), as well as communications connectors that exhibit suchimproved return loss performance. The techniques according toembodiments of the present invention may be particularly suitable foruse in Category 8 connectors that maintain backwards compatibility withthe Category 6a standard, as the amount of offending crosstalkcompensating crosstalk that is typically included in Category 6astandards-compliant connectors may appear as loads on the transmissionlines through the connectors that can make it difficult to maintain goodreturn loss performance at higher frequencies.

FIG. 18 is a flow chart that illustrates a method of designing atransmission line for a communications connector according to certainembodiments of the present invention. As shown in FIG. 18, operationsmay start at block 400 with a circuit designer designing a transmissionline of the connector (block 405). This transmission line model may takeinto account loads that are placed on the transmission line such ascrosstalk that is received from other transmission lines in theconnector, transmission lines in adjacent connectors, etc. Once aninitial design for the transmission line has been prepared, modelingtechniques may be used to simulate the return loss performance of thetransmission line, or alternatively, a test circuit may be built and thereturn loss performance may be measured (block 410). At block 415, adetermination is made as to whether or not the modeled or measuredreturn loss performance is acceptable. If it is, operations may then end(block 465). If not, operations may then continue to block 420, wherethe transmission line design is modified to include an impedancemismatch, the magnitude of an existing impedance mismatch is varied,and/or the delay between two impedance matches is varied.

Next, modeling techniques (or, alternatively, actual measurements) maybe used to simulate the return loss performance of the modifiedtransmission line design (block 425). At block 430, a determination isthen made as to whether or not the return loss performance of thetransmission line was improved in a desired fashion (e.g., within acertain frequency range) by the modifications of block 420. If so,operations may proceed to block 440 where the change that was made tothe transmission line design at block 420 is then increased further. Forexample, if at block 420, an impedance mismatch was increased by 5%,then at block 440, that same impedance mismatch may be increased byanother 5%. As another example, if at block 420, an impedance mismatchwas decreased by 5%, then at block 440, that same impedance mismatch maybe decreased by another 3%. Operations then proceed back to block 425where the return loss may be modeled again, and then operations proceedto block 420.

If, at block 430 it is determined that the return loss performance wasnot improved, then operations proceed to block 445 where the change thatwas made to the transmission line design at block 420 is then removedand instead a change to the impedance mismatch or delay is made in theopposite direction. For example, if at block 420, an impedance mismatchwas increased by 5%, then at block 445, the 5% increase in the impedancemismatch may be omitted and instead replaced with a 3% reduction in theimpedance mismatch. Operations may then proceed to block 455, where adetermination is made as to whether the return loss performance isacceptable. If it is, operations may end (block 465). If not, operationsmay return to block 420 and another impedance mismatch may be created ora different impedance mismatch or delay may be varied.

According to further embodiments of the present invention, the designmethod of FIG. 18 may be modified as follows. After designing thetransmission line at block 405, measurements or simulation may be usedto determine an approximate total value of the capacitive impairmentC_(imp) on the transmission line. This capacitive impairment value Cimpmay include both within pair coupling and coupling to other transmissionlines. Then at block 420, a box section filter may be created by addingbetween the conductors of the transmission line at issue a capacitorC_(in) that has a value of approximately C_(imp), and by further addingtwo inductors L+ and L−, each which have an initial value of1/(4*pi²*C_(imp)*f²), where f is the upper limit of the frequency rangeover which the return loss is to be improved. If this results inacceptable return loss performance, the method may then be terminated.If not, the values of one or more of C_(imp), L+ and/or L− may beadjusted and the return loss performance may be re-simulated untilacceptable return loss performance is achieved.

In some embodiments, communications connectors such as communicationsplugs (and patch cords that include such plugs) are provided that meetpre-determined performance criteria. For example, in some embodiments,communications plugs are provided that comply with the offendingpair-to-pair crosstalk specifications set forth in the Category 6astandard, which is incorporated herein by reference, while eachtransmission line through the plug also maintains an average return lossof at least 15 dB over the frequency range from 1.0 GHz to 2.0 GHz. Inother embodiments, these plugs can exhibit even higher return lossperformance while complying with the offending pair-to-pair crosstalkspecifications set forth in the Category 6a standard, such as havingeach transmission line through the plug maintain an average return lossof at least 20 dB over the frequency range from 1.0 GHz to 2.0 GHz orhaving each transmission line through the plug maintain an averagereturn loss of at least 18 dB over the frequency range from 500 MHz to1.5 GHz.

Pursuant to some embodiments of the present invention, the techniquesdiscussed above may also be used to improve the return loss performanceand/or the insertion loss performance of Cat 6a plugs. By way ofexample, FIG. 19 is a circuit diagram of a model of a conventional Cat6a test plug 400. As shown in FIG. 19, the plug 400 includes a total ofeight conductive paths 401-408 that are arranged a four differentialtransmission lines 411-414. The plug 400 includes a plurality of loadsincluding a first mutual inductance 421 where transmission lines 403 and404 inductively couple and a second mutual inductance 422 wheretransmission lines 405 and 406 inductively couple, as well as capacitivecouplings 423-426. The loads 421-426 may be provided so that the plugmeets the crosstalk requirements set forth in the Cat 6a standard.

FIG. 20 illustrates the simulated return loss of the plug 400 of FIG.19. In FIG. 20, curves 431-434 indicate the return loss on the fourtransmission lines 411-414, respectively, in the forward direction, andcurves 435-438 indicate the return loss on the four transmission lines411-414, respectively, in the reverse direction (note that curves 432and 434 overlap exactly and that curves 436 and 438 overlap exactly, sothat FIG. 20 appears to only have six curves as opposed to eightcurves). As shown in FIG. 20, the test plug 400 may provide goodperformance. However, the plug 400 may also be difficult to implement.As is also shown in FIG. 20, inadvertent resonance occurs on at leastsome of the differential pairs, including a very clear resonance on pair3 in the reverse direction (curve 437) which results in a local maximain the return loss on that pair. However, as is also apparent from FIG.20, the improvement in return loss that is provided by this inadvertentresonance occurs well outside the frequency range of interest, and hencedoes nothing to improve the performance of the plug 400.

Pursuant to embodiments of the present invention, Cat 6a plugs may beprovided that exhibit improved return loss performance. For example,FIG. 21 is a circuit diagram of a Cat 6a plug 500 that includes discreteelements that generate impedance mismatches that are tuned to improvereturn loss and insertion loss performance. As shown in FIG. 21, theplug 500 is similar to the plug 400 of FIG. 19 (note that elements521-526 of plug 500 correspond exactly to elements 421-426 of plug 400),but the plug 500 further includes additional discrete elements 541-548that create impedance mismatches that are tuned to provide enhancedreturn loss performance.

FIG. 22 illustrates the simulated return loss of the plug 500 of FIG.21. In FIG. 22, curves 531-534 indicate the return loss on the fourtransmission lines 511-514, respectively, in the forward direction, andcurves 535-538 indicate the return loss on the four transmission lines511-514, respectively, in the reverse direction (note that curves 532,534, 536 and 538 overlap exactly so that FIG. 22 appears to only havefive curves as opposed to eight curves). As shown in FIG. 22, the testplug 500 may provide excellent performance, and may providesignificantly better return loss performance as compared to the plug400.

FIG. 23 is a graph illustrating the simulated return loss performance ofthe transmission lines of another model of a Cat 6A plug that includesboth discrete elements and transmission line impedance imbalances thattogether provide return loss and insertion loss enhancements. In FIG.23, curves 631-634 indicate the return loss on the four transmissionlines through the plug in the forward direction, and curves 635-638indicate the return loss on the four transmission lines 511-514 throughthe plug in the reverse direction (note that curves 632, 634, 636 and638 overlap exactly so that FIG. 23 appears to only have five curves asopposed to eight curves). As shown in FIG. 23, this that also providesexcellent return loss performance. Additionally, as can be seen in FIG.23 a plurality of peaks are generated in the return loss spectra forsome of the pairs.

Pursuant to further embodiments of the present invention, methods ofimproving the return loss on a transmission line through acommunications connector are provided in which a plurality of impedancemismatches are intentionally provided along the transmission line. Theseimpedance mismatches may be caused by, for example, loads provided alongthe transmission line due to crosstalk circuits, discrete elements addedalong the transmission line to create impedance mismatches, and/or bychanges in the characteristics of the transmission line (e.g., changingthe width, thicknesses or spacing of the conductors thereof, changingthe dielectric constants of surrounding materials, changing therelationship of the conductors with respect to image planes, etc.).Then, the magnitude of one or more of the impedance mismatches, thedelay between one or more of the impedance mismatches and/or the numberof impedance mismatches provided may be adjusted in order to reduce arate of decrease in the return loss of the transmission line as afunction of increasing frequency within a desired frequency operatingrange of the transmission line.

In some embodiments, the transmission line may be a differentialtransmission line through a plug of a patch cord. In some embodiments,the rate of decrease in the return loss of the transmission line as afunction of increasing frequency may be reduced sufficiently such that alocal maxima is generated in the return loss spectra.

While embodiments of the present invention have been described aboveprimarily with respect to communications plugs, it will be appreciatedthat the techniques disclosed herein are equally applicable tocommunications jacks. FIGS. 24 and 25 illustrate a portion of a printedcircuit board of a communications jack according to further embodimentsof the present invention that includes electrical circuits in the formof inductors along a transmission line of the jack that createpre-selected impedance mismatches that are used to improve the returnloss and/or the insertion loss of the transmission line over a desiredfrequency range. As discussed above, inductors that are used to createsuch impedance mismatches may be implemented by, for example, changingthe width, thickness or spacing of the conductive traces/elements on theprinted circuit board that form a transmission line segment or byincluding discrete inductors along the transmission line. In thisparticular embodiment, the inductors are implemented as discretecomponents in the form of solenoid inductors.

In particular, FIG. 24 is a top view of a portion of a multi-layerprinted circuit board 710 of an RJ-45 communications jack 700. A pair ofprinted circuit board based solenoid inductors 720, 740 are implementedwithin the printed circuit board 710. FIG. 25 is an oblique view of asmaller portion of printed circuit board 710 that is illustrated in FIG.24 that shows the solenoid inductors 720, 740 in greater detail. In theparticular embodiment of FIGS. 24-25, the solenoid inductors 720, 740are implemented on the respective conductive paths through the printedcircuit board 710 for pair 3 (see FIG. 2) of the RJ-45 jack. Eachsolenoid inductor 720, 740 is implemented as a single-turn solenoid,although it will be appreciated that additional turns could be includedin other embodiments if larger series inductance values are needed.

As shown in FIGS. 24-25, the printed circuit board 710 includes aplurality of metal-plated holes 712 and a plurality of metal-platedholes 714. The metal-plated holes 712 receive the eight jackwirecontacts of the RJ-45 jack 700. Only seven of the eight metal-platedholes 712 (namely holes 712-1 through 712-7) are visible in the portionof the printed circuit board 710 shown in FIG. 24. The metal-platedholes 714 receive the eight insulation displacement contacts of theRJ-45 jack 700. Only two of the eight metal-plated holes 714 (namelyholes 714-3 and 714-6) are visible in the portion of the printed circuitboard 710 shown in FIG. 24. In FIG. 25, the dielectric material of theprinted circuit board 710 is made translucent to illustrate conductivestructures formed on layers of the printed circuit board 710 that arebelow the top layer, and only the conductive structures that are part ofpair 3 are shown to simply FIG. 25.

Solenoid inductor 720 is formed in printed circuit board 710 using aplurality of conductive traces 722, 724, 726 and several conductive vias730, 732, 734 that extend vertically through the printed circuit board710. The conductive traces 722, 724, 726 are relatively wide as they aresignal current carrying traces that carry higher current levels.Conductive via 730 physically and electrically connects trace segment722 to trace segment 724. Conductive via 732 physically and electricallyconnects trace segment 724 to trace segment 726. Conductive via 734physically and electrically connects trace segment 726 to another tracesegment 728 that connects (either directly or indirectly) tometal-plated hole 712-6 that receives a jackwire contact. The end ofconductive trace 722 opposite conductive via 730 connects to themetal-plated insulation displacement contact hole 714-6. Themetal-plated via 714-6 is not deliberately implemented to be part of thesolenoid inductor 720, but because of the size of this structure and itslocation immediately adjacent the solenoid inductor 720 it will affectthe inductances and hence the impact of the metal-plated hole 714-6 isaccounted for when tuning the solenoid inductor 720.

Similarly, the printed circuit board based solenoid inductor 740 isformed in printed circuit board 710 using a plurality of conductivetraces 742, 744, 746 and two conductive vias 750, 752 that extendvertically through the printed circuit board 710. Conductive via 750physically and electrically connects trace segment 742 to trace segment744. Conductive via 752 physically and electrically connects tracesegment 744 to trace segment 746. Only two conductive vias 750, 752 areused to implement solenoid inductor 740 as conductive trace 746continues on the same layer of printed circuit board 710. Conductivetrace segment 746 connects (either directly or indirectly) tometal-plated hole 712-3 that receives a jackwire contact. The end ofconductive trace 742 opposite conductive via 750 connects tometal-plated insulation displacement contact hole 714-3. Themetal-plated hole 714-3 is not implemented to be part of the solenoidinductor 740, but again because of its size and location immediatelyadjacent the solenoid inductor 740 it will affect the inductances, andhence the impact of the metal-plated hole 714-3 is accounted for whentuning the solenoid inductor 740.

Solenoid inductor 720 acts to increase the impedance of the conductivepath on printed circuit board 710 that connects metal-plated insulationdisplacement contact hole 714-6 to the metal-plated jackwire contacthole 712-6. Solenoid inductor 740 acts to increase the impedance of theconductive path on printed circuit board 710 that connects metal-platedinsulation displacement contact hole 714-3 to the metal-plated jackwirecontact hole 712-3. As a result, the impedance of the differentialtransmission line for pair 3 is increased. In an RJ-45 jack, return lossmay be higher than desired because the impedance of the differentialtransmission line for pair 3 is below 100 ohms, and hence the solenoidinductors 720, 740 may increase the impedance of the differentialtransmission line for pair 3 and hence improve the return lossperformance of the jack 700.

The use of solenoid inductors 720, 74Q may be advantageous becausespirals of the needed inductance and current capacity may be too largeto fit in the available space on the printed circuit board 710 for anRJ-45 jack. Additionally, the magnetic fields of solenoid inductors 720,740 may be nearly perpendicular to those of other circuits such ascrosstalk compensation circuits that are implemented on the printedcircuit board 710, and hence have less unintended parasitic couplingwith such circuits.

The inductance of an isolated solenoid inductor is determined by thearea of the loop current, the number of loops, and the effectivemagnetic permeability (mu) of the volume threaded by the magnetic field.Here, the magnetic permeability is equal to one because no ferrites orother such materials are present in the printed circuit board 710, sothe inductance is determined entirely by the geometry of the conductivetraces and vias. In the embodiments of FIGS. 24-25, the two solenoidinductors 720, 740 are located close enough to one another so that theycouple with each other (the magnetic field from each solenoid inductor720, 740 partially threads through the other), creating a mutualinductance which affects the impedance of two solenoid inductors 720,740 when considered/used as a differential pair.

The necessary inductance is achieved by adjusting the number of loops(here each solenoid inductor 720, 740 has a single loop), the lengths ofthe loops (their height is fixed by the thickness of the printed circuitboard 710), and the handedness or sense of the turns of each solenoidinductor 720, 740 (which controls whether the magnetic fields of eachcircuit will tend to add together and reinforce each other, or opposeeach other). Here it is beneficial (more effective use of the space) tohave the loops turn in opposite senses geometrically, which results intheir magnetic fields reinforcing each other because the instantaneouscurrents in each circuit flow in opposite directions (they are membersof a differential pair, so instantaneous current is flowing out of oneinsulation displacement contact while flowing into the other insulationdisplacement contact).

The solenoid inductors in the depicted embodiment are located near themetal-plated insulation displacement contact holes 714-3, 714-6. Inother embodiments, the solenoid inductors would not necessarily be closeto the insulation displacement contacts holes 714.

While the solenoid inductors 720, 740 each include a single loop, itwill be appreciated that in other embodiments additional loops can beadded by adding additional conductive trace segments and conductive viashaving the same configuration immediately adjacent the first loop aspart of the conductive path.

It will also be appreciated that the solenoid inductors that aredisclosed herein may be used in other applications where it is desirableto implement a series inductance along a transmission line including,for example, other types of return loss circuits such as those disclosedin U.S. Pat. No. 7,326,089 and/or in applications where seriesinductances are used in implementing frequency dependent crosstalkcompensation circuits as disclosed, for example, in U.S. Pat. No.7,190,594. The entire contents of U.S. Pat. No. 7,326,089 and U.S. Pat.No. 7,190,594 are incorporated herein by reference. It will further beappreciated that the above-disclosed solenoid inductors may beimplemented in communications plugs as well in further embodiments.

While the above embodiments have focused on connectors that includeprinted circuit boards, it will be appreciated that the techniquesaccording to embodiments of the present invention may be implemented inconnectors that do not include printed circuit boards such as plugs orjacks that use lead frame implementations.

While the above embodiments have focused on connectors that includedifferential transmission lines, it will be appreciated that thetechniques according to embodiments of the present invention may also beimplemented in connectors that use single-ended transmission lines. Forexample, in further embodiments, plugs may be provided in which one ormore of the differential pairs are implemented as two single-endedtransmission lines that have very low coupling with each other (and withthe conductors of the remaining pairs) as opposed to using a singledifferential transmission line.

The present invention is not limited to the illustrated embodimentsdiscussed above; rather, these embodiments are intended to fully andcompletely disclose the invention to those skilled in this art. In thedrawings, like numbers refer to like elements throughout. Thicknessesand dimensions of some components may be exaggerated for clarity.

Spatially relative terms, such as “top,” “bottom,” “side,” “upper,”“lower” and the like, may be used herein for ease of description todescribe one element or feature's relationship to another element(s) orfeature(s) as illustrated in the figures. It will be understood that thespatially relative terms are intended to encompass differentorientations of the device in use or operation in addition to theorientation depicted in the figures. For example, if the device in thefigures is turned over, elements described as “under” or “beneath” otherelements or features would then be oriented “over” the other elements orfeatures. Thus, the exemplary term “under” can encompass both anorientation of over and under. The device may be otherwise oriented(rotated 90 degrees or at other orientations) and the spatially relativedescriptors used herein interpreted accordingly.

The present invention is directed to communications connectors such asRJ-45 plugs. As used herein, the terms “forward” and “front” andderivatives thereof refer to the direction defined by a vector extendingfrom the center of the plug toward the portion of the plug that is firstreceived within a plug aperture of a jack when the plug is mated with ajack. Conversely, the term “rearward” and derivatives thereof refer tothe direction directly opposite the forward direction.

Well-known functions or constructions may not be described in detail forbrevity and/or clarity. As used herein the expression “and/or” includesany and all combinations of one or more of the associated listed items.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the invention. Asused herein, the singular forms “a”, “an” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises”,“comprising”, “includes” and/or “including” when used in thisspecification, specify the presence of stated features, integers, steps,operations, elements, and/or components, but do not preclude thepresence or addition of one or more other features, integers, steps,operations, elements, components, and/or groups thereof.

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this invention belongs. It will befurther understood that terms, such as those defined in commonly useddictionaries, should be interpreted as having a meaning that isconsistent with their meaning in the context of the relevant art andwill not be interpreted in an idealized or overly formal sense unlessexpressly so defined herein.

The foregoing is illustrative of the present invention and is not to beconstrued as limiting thereof. Although exemplary embodiments of thisinvention have been described, those skilled in the art will readilyappreciate that many modifications are possible in the exemplaryembodiments without materially departing from the novel teachings andadvantages of this invention. Accordingly, all such modifications areintended to be included within the scope of this invention as defined inthe claims. The invention is defined by the following claims, withequivalents of the claims to be included therein.

1-5. (canceled)
 6. A method of improving the return loss on adifferential transmission line through a communications connector, themethod comprising: dividing the differential transmission line into atleast a first segment, a second segment and a third segment, wherein afirst impedance mismatch between the impedances of the first and secondsegments differs by at least 20%, and a second impedance mismatchbetween the impedances of the second and third segments differs by atleast 20%.
 7. The method of claim 6, further comprising selecting thefirst impedance mismatch and the second impedance mismatch to improvethe return loss performance of the differential transmission line in apre-selected frequency range.
 8. The method of claim 7, furthercomprising selecting a length of the first segment, a length of thesecond segment and a length of the third segment to improve the returnloss performance of the differential transmission line in a pre-selectedfrequency range.
 9. The method of claim 6, wherein the first segment ofthe differential transmission line comprises conductive traces on aprinted circuit board and the third segment comprises a pair of plugcontacts.
 10. The method of claim 6, wherein the first impedancemismatch is formed at least in part by implementing the first segment ofthe differential transmission line using conductive traces on a printedcircuit board having first widths and/or a first spacing and byimplementing the second segment of the differential transmission lineusing conductive traces on the printed circuit board having secondwidths and/or a second spacing that differ from the respective firstwidths or first spacing.
 11. The method of claim 6, wherein at least oneof the first impedance mismatch or the second impedance mismatch isgenerated at least in part by a discrete reactance.
 12. The method ofclaim 6, wherein at least one of the first impedance mismatch or thesecond impedance mismatch is generated at least in part by providing abox-section filter along the differential transmission line.
 13. Amethod of improving the return loss on a transmission line through acommunications connector, the method comprising: including a pluralityof impedance mismatches along the transmission line; adjusting themagnitude of one or more of the impedance mismatches, the delay betweenone or more of the impedance mismatches and/or the number of impedancemismatches provided in order to reduce a rate of decrease in the returnloss of the transmission line as a function of increasing frequencywithin a desired frequency operating range of the transmission line. 14.The method of claim 13, wherein the transmission line comprises adifferential transmission line through a plug of a patch cord.
 15. Themethod of claim 13, wherein rate of decrease in the return loss of thetransmission line as a function of increasing frequency is reducedsufficiently such that a local maxima is generated in the return lossspectra. 16-18. (canceled)